1. Field of the Invention
The invention relates to driver circuits, and in particular to features of driver circuits directed to the control of harmonic content, common-mode noise, and reflective losses in the driver output. More particularly, the invention relates to controlling transient aberrations and reflections in data pulses transmitted by a current-switch/driver to a data-transmission line. Even more particularly, the invention relates to minimizing transient aberrations and signal reflections as part of and in addition to the minimization of electromagnetic-interference (EMI) arising from high-frequency (&gt;50 Mhz) data transmission over unshielded metal conductors.
2. Prior Art
Great advantages attach to high-frequency data transmission and in particular to such transmission on unshielded transmission lines such as the twisted pairs of copper wires often used to couple local area computer networks. Unfortunately, the high-frequency pulses making up the data give rise to unacceptable levels of EMI and also suffer impedance-mismatch losses--problems which worsen as the transmission frequency increases. Major sources of EMI are: (a) common-mode noise on the transmission line arising from (i) capacitive coupling of the transmitter output to its input and intermediate nodes and (ii) noise on the power rail forming the common reference for the output signals, (b) ringing and overshoot (collectively, "transient aberrations") of the pulses, and (c) asymmetries between the pulses' rising and falling edges. Generally stated, the condition for having no EMI from the transmission line requires that the line not give rise to any alternating (ac) electromagnetic field at a distance. Since the transmission line is supposed to carry both a signal and the complement of that signal, in theory the problem does not exist; the field at a distance arising from the signal should be equal in magnitude and opposite in phase to the field at a distance arising from the signal complement and hence the superposition of these two fields should lead to an ac nullity. Stated differently, at a distance from a twisted pair of conductors carrying a signal and a signal complement, respectively, the ac electromagnetic field of the two conductors will appear to be zero because of cancellation of two equal-magnitude/opposite-phase ac signals. Now, anything that breaks the signal/signal-complement anti-symmetry will lead to a non-vanishing ac electromagnetic field at a distance and hence to EMI at a frequency comparable to the signal frequency. This occurs, for example, if both signals are referenced to a source that is itself time-varying. It also occurs if, through capacitive coupling or other means, a time-varying signal is fed onto one or both conductors of the transmission line pair. It occurs in an especially detrimental fashion if there is ringing and/or overshoot at the pulse edges. (These transient aberrations cannot cancel out and they provide a net signal which is both of high amplitude and of extremely high frequency.) Non-vanishing ac fields-at-a-distance also occur if the falling edge of the pulse is not symmetric with the rising edge. E.g., if the rising edge is steeper than the falling edge, then the rising pulse on the signal line added to the corresponding falling pulse on the complement signal line yields a non-zero signal which varies with time. These various signal problems giving rise to EMI are inherent in the prior-art drivers; attempts to minimize them generally have been limited to introducing LRC filters on the transmission line near where it couples with the transmitter and receiver. (In addition, choke coils and isolator transformers typically are used at both ends of the transmission line.)
Unfortunately, the LRC filters introduced in the prior art to address the EMI and impedance mismatch problems associated with high-frequency transmission introduce problems of their own. These include but are not limited to signal dispersion and ringing. Because the LRC filters employed with the prior art are of limited band width, they are unable to adequately reduce EMI and reflective losses across the entire signal bandwidth no matter how well they are tuned to do the job at the fundamental carrier frequency. Frequencies of interest at the present level of development are on the order of 62.5 MHz; rise and fall times of the pulses making up the data trains of interest are on the order of 600 picoseconds. It is this very fast rise/fall time that leads to the EMI-generating transient aberrations. The LRC filters are particularly ineffective in addressing the transient aberration problems at these high frequencies and can in fact exacerbate them. A final consideration is that circuitry conveying pulses with extremely fast rise- and fall-times is very unforgiving--in the sense that a junction adequate for lower frequencies can be the site of significant ringing at the high frequencies implicit in such short rise/fall times. A high premium attaches to the care with which intra-circuit connections are made if the circuit is going to be carrying such signals. This need can be relaxed somewhat if the rise/fall time can be lengthened (within the constraints implicit in the underlying transmission frequency, on the order of 62.5 MHz and higher).
FIG. 1 shows a typical prior-art system incorporating an ECL switch/driver as the transmitter--with output V.sub.o and complementary output V.sub.o B. (The inset FIG. 1a shows the ECL switch/driver in detail.) LRC filters--F1 and F2--are placed on the transmission line to reduce the EMI and to provide impedance matching. In addition, choke coils T2 and T4 are shown, along with isolation transformers T1 and T3. The choke coils are used to reduce common-mode noise. It can be seen from the ECL circuit constituting the switch/driver that any noise picked up by the high-potential power rail V.sub.cc will be imposed on both members of the transmission-line pair, V.sub.o and V.sub.o B. In general this will be high-frequency noise and so will contribute to the EMI problem. It would be preferable to have the output and the output complement referenced to the low-potential power rail GND rather than to V.sub.cc ; it is the nature of the GND rail to be more tightly coupled to the external reference and consequently less susceptible to pick-up than is the V.sub.cc rail. It is, however, in the nature of the ECL drivers that the output signals are referenced to V.sub.cc.
By shifting to a MOS-based switch/driver, it is possible to generate current pulses referenced to the low-potential power rail GND. FIG. 2a depicts a prior-art MOS switch/driver analogous to the ECL circuit of FIG. 1. The current I.sub.M to be switched between the two branches is fixed by some known means represented by the current regulator/generator symbol in the line running from the high-potential power rail V.sub.cc to the common source nodes of the PMOS output transistors QA and QB. The output transistors QA and QB are switched by the CMOS inverter stages coupled between V.sub.cc and GND and controlled by the input and complementary input signals, E and EB, respectively. These are standard CMOS inverters with the relative sizes of the PMOS and NMOS devices skewed so as to provide symmetric output signals to the gates of QA and QB in response to symmetric input signals at E and EB, respectively. (The skewing involves making the PMOS channel wider than the NMOS channel. The degree to which this must be done depends on the particular process used to fabricate the device; for a typical prior art process, symmetric CMOS input signals generate symmetric CMOS output signals if the P-Channel is three times the width of the N-Channel.) With this arrangement, the control gate of each driver transistor is pulled to either V.sub.cc --which cuts the driver transistor completely off--or to GND, which turns it fully on. Thus, the entire current I.sub.M either passes through QA and to ground through the resistor coupling the QA drain to GND--making V.sub.o =I.sub.M (R.sub.o /2) and V.sub.o B=GND--or it passes through QB to GND resulting in the opposite signals to the transmission lines. This circuit has the advantage from the standpoint of common-mode noise reduction that V.sub.o and V.sub.o B are referenced to the more stable power rail, GND.
In addition to V.sub.cc fluctuation, another major source of common-mode noise is capacitive coupling of the input signal to other nodes of the switch/driver. With reference to FIG. 1, such coupling between the base nodes of transistors QX and QY, respectively, and the outputs V.sub.o and V.sub.o B will generate common-mode noise of the same high frequency as the input. Regardless of the particular driver circuit involved, there is a certain minimum input/output capacitive coupling which cannot be eliminated. Once that level is approached, efforts at further reduction in this component of common-mode noise must look to minimizing the voltage swing of the input signals themselves. With the ECL switch circuit depicted in FIG. 1, the minimum swing must be enough to turn the npn transistors on and off completely. I.e., for a good on/off ratio at the driver output, the constant tail current I.sub.f must pass completely through QX or completely through QY. This requires a minimum input swing at E and EB of about 0.3 V, though the swing used in actual practice is more like 0.8 V.
With respect to minimization of the input swing, the MOS-based driver is not as good as the ECL unit; frequently a full rail-to-rail swing is used for switching MOS transistors. This can be seen to be the case with the circuit of FIG. 2a. It can also be seen from FIG. 2a that it is overkill to use a full rail-to-rail voltage swing to switch QA and QB on and off. All that is really necessary for QA shut-off is that its gate be at a voltage greater than V.sub.s -V.sub.T, where V.sub.s is the common source voltage in this circuit and V.sub.T --is the threshold source--gate voltage for turning on the MOS transistor QA. The specific value of the common source voltage depends on the exact nature of the circuit sourcing I.sub.M and the gain of the member of the QA, QB pair which is conducting. In any event, this means that biasing the gate of QA, for example, to the common source voltage--i.e., making V.sub.gs =0 for QA--will certainly ensure that QA is non-conducting. Similar remarks follow for QB.
FIG. 2b shows the prior-art switch/driver of FIG. 2a with a particular current-sourcing mechanism selected, namely a PMOS transistor Q7 operating in its saturation mode and with its gate biased by an off-chip-generated V.sub.BIAS so as to establish by known means a mirrored current I.sub.M independent of operating temperature, power supply voltage, and vagaries in the process by which the chip was manufactured. With this particular means for regulating the current to be switched, it is possible and desirable to interpose PMOS transistors (QC and QD) between the CMOS stages and V.sub.cc. This ensures that the "turn-off" voltage applied to the QA and QB control nodes makes V.sub.gs approximately zero, and that the voltage swing is reduced by the drop across QC (or QD) when it is conducting. This reduction in input swing voltage significantly reduces the high frequency noise capacitively coupled onto the transmission line. (The ringing and overshoot problems, however, still remain, since they are associated with the fast rise/fall time, which exists for the MOS-based circuit as for the ECL circuit.)
Another problem needs to be overcome in order to reap the common-mode noise advantages of the MOS-based switch/driver; it relates to signal shape and the complicated way in which the conduction current of a MOS transistor varies in response to the voltage applied at its gate node, particularly in contrast to the simple response curve of bipolar transistors. That is, unlike the simple transfer function--Ktanh(V.sub.IN /V.sub.T)--describing the switching of the bipolar pair of FIG. 1 as a function of control node voltage V.sub.IN, the transfer function describing the MOS differential pair is quite complicated, involving as it does several modes of transistor operation. For example, the "turning-on" device may be in its saturation region while the "turning-off" device is in its subthreshold--weak inversion--region. This means that the incremental current-changes in the MOS device turning on will not match in magnitude the simultaneous current changes in the device turning off. The result is that for symmetric input pulses at E and EB (and consequently, by design, symmetric pulses at the control nodes of QA and QB in accordance with FIG. 2a and FIG. 2b ), the output current pulses through the resistors Ro/2 in the circuits depicted in FIG. 2a and FIG. 2b will not be symmetric; in particular, the rising edge of the turning-on current pulse will be qualitatively different from the falling edge of the turning-off current pulse. This is a problem which must be resolved in any MOS-based switch/driver to be used in a situation where EMI is a concern.
As noted above, going to the MOS-based switch/driver does not resolve the transient aberration problem. It arises in both the ECL-based and MOS-based prior art from the extremely short switching time of these circuits--a switching time, as indicated, on the order of 0.6 nsec. Fortunately, it is only necessary to lengthen this time by a factor of three or so to practically eliminate the transient aberrations. (Since, at the transmission frequencies in question--about 62.5 MHz--the pulse length itself is on the order of 8 nsec, lengthening the rise/fall time to 2 nsec leaves a sufficient "eye" for the pulses to be processed by the receiver.) Unfortunately, lengthening the rise/fall time turns out to be a fix more easily prescribed than done. For example, extending the rise/fall time by placing an RC time constant (capacitive low-pass filter) at the output of the switch/driver is unacceptable because of the resulting changes in the transmitter's output impedance--a serious concern because of the reflective losses which result from impedance mismatch at these frequencies.
Therefore, what is needed is a high-frequency switch/driver which generates current pulses referenced to the low-potential power rail, a switch/driver that can transmit current pulses symmetric in their rising and falling edges in response to symmetric input voltages at E and EB. What is also needed is that such a switch/driver be able to produce output pulses with a good "on" to "off" ratio in response to a minimal input voltage swing. Finally, what is needed is a means of increasing the rise- and fall-times of the generated pulses without degrading the output impedance of the switch/drivers producing the pulses.